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1
All-planar Gain-enhanced and Minimized Patch Antenna
Student: Chang-Yi Tsai
Institude of Communication Engineering
National Chiao Tung University
ABSTRACT
Utilizing the recently developed Electric-Magnetic-Electric (EME) composite
Advisor: Dr. Ching-Kuang C. Tzuang
metal strips, which can increase the slow-wave factor and have a broad stopband, two applications are introduced in this thesis. First one uses slow-wave structures to minimize the patch antenna employing PCB (printed-circuit-board) fabrication process. The resonant length and area of the slow-wave patch antenna are reduced to be 76 % and 58 % when compared with the conventional one. Gain enhancement of the patch antenna by suppressing surface waves is the other application in our studies. According to the experimental results, the peak gain of the design patch antenna is 1.03 dBi higher than the conventional one. Statements and measurement results of both patch antennas are also included in this thesis.
2
Table of Contents
Abstract (Chinese)…………..……………………………….……..……i
Abstract (English)…………….…………………………………..…….ii
Acknowledgments……………………………………………………...iii
Table of Contents…………………………………………………….…iv
List of Figures…………………………………………………………..vi
List of Tables…………………………………………….………………x
CHAPTER 1 Introduction………..………………………………….1
1.1 Introduction…………………………………………….1 1.2 Organization of This Thesis……………………….…...3
CHAPTER 2 Guiding Characteristics of Periodic Structure..….....4
2.1 Introduction……………………………….……………4
2.2 Electric-Magnetic-Electric (EME) Microstrip…….….10 2.3 Guiding Characteristics of Electric-Magnetic-Electric
(EME) Microstrip……………………..………………12
CHAPTER 3 Size Reduction of Microstrip Patch Antennas Using
Periodic Structures…………………………….....….17 3.1 Introduction…………………………………………...17
3
3.2 Patch Antenna Design……………..……………….....22
3.3 Statements of Slow-wave Microstrip Patch Antenna…27 3.4 Measurement Results…………………………………32
CHAPTER 4 Patch Antenna Gain Enhancement Utilizing
Periodical Structure…………………………………36 4.1 Introduction…………………………………………...36
4.2 Conventional Patch Antenna Design…...………….....38 4.3 Periodical Structure Design…….……..……………….…..39 4.4 Fabrication and Experiments………………………….42 4.5 Measurement Results…………………………………44
CHAPTER 5. Conclusions.………………………..………….……..50
References...…………………………………………………………….51
4
List of Figures
Figure 2.1 A PBG structure for microstrip lines based on etching a 2D
Figure 2.2 Figure 2.2 Figure 2.3 Figure 2.4 Figure 2.5 Figure 2.6
lattice of circles in the ground plane. This PBG ground plane can be used for second harmonic tuning of a power amplifier………………………………………….…………7
(a) Cross section of a high-impedance surface, fabricated as
a printed circuit board. The structure consists of a lattice of metal plates, connected to a solid metal sheet by vertical conducting vias…………………………………...….….….8
(b) Top view of the high-impedance surfaces, showing a
triangular lattice of hexagonal metal plates.……………..…8
Schematic of the compact photonic–bandgap (UC-PBG)
structure. A microstrip line is on the top side of the substrate while the connected-pads is on the other side.……………...9
Schematic of the microstrip line with metal strip replaced by
electric-magnetic-electric (EME) surfaces.………………..11
Photograph of the 15-cell long Electric-Magnetic-Electric
(EME) microstrip..…………………………………...……14
Measured and computed scattering parameters for the EME
5
microstrip………………………………………………….15
Figure 2.7 Slow-wave factor and characteristic impedance of the MS
microstrip (or EME microstrip of 100% filling) in comparison with the uniform microstrip.………………….16
Figure 3.1 Configuration of microstrip antenna of very high
Figure 3.2 Figure 3.3 Figure 3.4 Figure 3.5 Figure 3.5 Figure 3.5 Figure 3.6 Figure 3.7 Figure 3.8
permittivity………………………………………………...19
Schematic of probe-fed patches incorporating single shorting
posts. (a) Circular. (b) Rectangular………………………..20
Geometry of rectangular microstrip antenna with
chip-resistor loading……………………………………….21
Patch with Slow-wave (SWF) structure…………………...21 (a) The geometry of a rectangular patch microstrip
antenna…………………………………………………….25
(b) Side view showing the electric fields……….……..….25 (c) Top view of the rectangular patch antenna…………...26 Microstrip edge feed with a quarter-wave transformer……26 Simulated scattering parameters for the EME microstrip with
15 cells…………………………………………………….29
Geometry of the EME patch with£`r = 3.38, h1 = 8mil, h2=
6
20mil……………………………………………...……….30
Figure 3.9 Microstrip edge feed with a quarter-wave transformer
(a) EME patch (b) Conventional patch..…………...……...31
Figure 3.10 Measured return loss of the conventional and slow-wave
patch antennas.………….…………………..……………..33
Figure 3.11 Measured radiation patterns for the slow-wave and
conference patch antennas in E-plane….…………….....…34
Figure 3.12 Measured radiation patterns for the slow-wave and
conference patch antennas in H-plane…………..………...35
Figure 4.1 Schematic of the microstrip patch antenna surrounded by
PBG lattices…………………………………………….....37
Figure 4.2 Microstrip edge feed with inset……………………..…..…38
Figure 4.3 The schematic of the microstrip antenna surrounded by EME
structures with£`r=3.38, h1=8mil, h2=20mil, d=56mil…..40
Figure 4.4 Computed scattering parameters for the design periodical
structure……………………………………………………41
Figure 4.5 (a) The conventional patch antenna …………………...…43 Figure 4.5 (b) The patch antenna surrounded by periodical
structures…………………………………………………..43
7
Figure 4.6 Measured return loss of the conventional and design patch
antennas……………………………………………………47
Figure 4.7 Measured normalized E-plane radiation patterns of the
design and conference patch antennas………………….…48
Figure 4.8 Measured normalized H-plane radiation patterns of the
design and conference patch antennas…………………….49
8
List of Tables
Table 4.1
Summary of two patch antennas. …………………………46
9
CHAPTER 1 Introduction
1.1 Introduction
Microstrip antennas are widely used in a broad range of many applications because of their advantageous features in terms of lightweight, low profile, low cost, and easy of integration with RF devices. However, the microstrip antennas size becomes larger for applications at low frequency. One such scenario is for use on a mobile communications handset. Frequency of less than 3 GHz requires conventional patches too large to be placed in a mobile handset.
Recently, several techniques have been proposed to minimize the microstrip patch antennas. Using shorting or resistive posts to reduce the patch size is very popular [1-3]. Although these methods can achieve considerable size reduction of patch antennas, the performances of gain and bandwidth will decrease. Another method to reduce the size is using high dielectric constant material [4]. However, this way usually results in narrow bandwidth and degradation of gain. Also, the limited availability of low lost and low cost, high dielectric constant material has another problem with this technique.
Previously, photonic band-gap (PBG) materials have been proposed to improve the performances of microwave circuits and antennas. The photonic band-gap
10
structure has been successfully used to improve the performances of microstrip antennas [5-6]. Here we use a novel technique for size reduction of microstrip patches incorporating Electric-Magnetic-Electric (EME) composite metal strips [7]. This newly developed periodic structure results in the increase of slow-wave factor (£f0/£fg), higher characteristic impedance and the same Q-factor with a microstrip line. Based on this result, size reduction of microstrip antennas can be achieved by incorporating EME metal strips at operative frequency.
Besides, the EME microstrip will exist a stopband due to the periodical structures. In the stopband, the periodical structures turn into a high-impedance state known as a magnetic surface. Surface waves will be suppressed in the stopband and power loss due to surface waves can be reduced. Because surface waves excited in the dielectric substrate will be suppressed, improvements of the microstrip antenna performances will be expected.
11
1.2 Organization of This Thesis
In chapter 2, we first introduce some periodic structures and their applications in microwave circuits and antennas. A short description and guiding characteristics of Electric-Magnetic-Electric (EME) composite metal strips are shown in this chapter. At the end of chapter 2, we show both theoretical and experimental results of EME microstrips when compared with conventional ones.
Chapter 3 shows how we use slow-wave character of the EME microstrip to minimize microstrip patch antennas. Some basic concepts are introduced to design a conventional microstrip patch antenna. Statements and measurement results of both conventional and slow-wave patch antennas are described in this chapter.
Chapter 4 describes gain enhancement of microstrip patch antennas utilizing periodic structures. By suppressing surface waves, the gain enhancement of a microstrip antenna is achieved when compared with the conventional one.
Chapter 5 concludes this thesis.
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CHAPTER 2
Guiding Characteristics of Periodic Structure
2.1 Introduction
Periodic structures for electromagnetic waves have been studied since the early days of microwave radars. One-dimensional periodic slots cut along a metallic waveguide, both resonant and non-resonant, have been used to actualize slotted-waveguide linear array antennas [8]. Two-dimensional periodic structures have found wide applications in frequency selective surface (FSS) and polarization diplexer designs [9]. With progress in fabrication technology in modern nano-fabrication technology, it is now possible to design and fabricate three-dimensional periodic structures, widely known as photonic bandgap (PBG) crystals, for important applications such as high-efficiency LEDs and nano-cavity lasers [10].
PBG structures have been studied in the optic region and have been applied to microwave and millimeter-wave circuits, recently [11-12]. This substrate is periodically loaded to create a so-called electromagnetic crystal whose surface-wave dispersion diagram presents a forbidden frequency range around the antenna’s operative frequency [13]. Since surface waves cannot propagate along the substrate, an increased amount of radiated power couples to space waves.
Several configurations and applications have been proposed to realize PBG
13
substrates. The first attempts were made by drilling a periodic pattern of holes in the substrate [14] or by etching a periodic pattern of circles in the ground plane [15]. A more effective approach, which makes use of a lattice of metallic pads connected to ground with vias [16-17]. Figure 2.1 shows a 2D photonic bandgap structure based on etching a 2D lattice of circles in the ground plane [15]. This PBG substrate with a 2D pattern of circles etched in the ground can be used for broadband second harmonic tuning of a power amplifier [18].
Figure 2.2 shows the periodic structure with triangular lattices of hexagonal metal plates [19]. This type of metallic electromagnetic structure can be characterized by having high surface impedance. Although it is made of continuous metal and conducts dc currents, it does not conduct ac currents within a forbidden frequency band. Applying this structure, antennas have advantage of the suppression of surface waves. An antenna on this type of structure produces a smoother radiation shape than a similar one on a conventional metal ground plane.
Another feature of the PBG structure is the realization of a slow-wave microstrip line. Slow-wave-mode propagation is generally used in reducing the dimension of distributed components in integrated circuits. A low-loss slow-wave microstrip have been demonstrated using a periodic structure named uniplanar compact photonic–bandgap (UC-PBG) [20]. Figure 2.3 shows the schematic of the
14
uniplanar compact photonic–bandgap (UC-PBG) structure. This periodic structure is realized with metal pads etched in the ground plane connected by thin lines to form a distributed LC network. This UC-PBG structure indicates significant increase in the slow-wave factor (£f0/£fg) than that of conventional ones over a wide frequency range. Many applications of UC-PBG substrates are proposed due to the unique character of low loss and slow-wave effect [6]. Microstrip bandpass filters with intrinsic spurious suppression were proposed to have better performances. Power amplifier of efficiency greater than 60% was reported by incorporating the stopband characteristic of such periodical microstrip structure [21]. Increasing new applications using this microstrip technology are proposed [22-23].
15
Figure 2.1 A PBG structure for microstrip lines based on etching a 2D lattice of circles in the ground plane. This PBG ground plane can be used for second harmonic tuning of a power amplifier.
16
Figure 2.2 (a) Cross section of a high-impedance surface, fabricated as a printed circuit board. The structure consists of a lattice of metal plates, connected to a solid metal sheet by vertical conducting vias. (b) Top view of the high-impedance surfaces, showing a triangular lattice of hexagonal metal plates.
17
Figure 2.3 Schematic of the compact photonic–bandgap (UC-PBG) structure. A microstrip line is on the top side of the substrate while the connected-pads is on the other side.
18
2.2 Electric-Magnetic-Electric (EME) Microstrip
This thesis uses the recently developed low-loss, slow-wave Electric-Magnetic-Electric (EME) microstrips [7]. The new microstrip replaces the conventional metal strip by the electric and magnetic surfaces. The conventional metal strip is replaced by adding a periodic structure alongside the metal strip(s) in the axial direction as shown in Figure 2.4. A number of coupled, connected metallic coils realize magnetic surfaces. These coils form a periodic array at the central plane of the microstrip. This Electric-Magnetic-Electric (EME) microstrip can be fabricated by using PCB (printed-circuit-board) fabrication process.
Like other PBG structures, stopband exists at certain frequency band because of the magnetic surface. Theoretical and experimental results are presented in next section.
19
Figure 2.4 Schematic of the microstrip line with metal strip replaced by electric-magnetic-electric (EME) surfaces.
20
2.3 Guiding Characteristics of Electric-Magnetic-Electric (EME)
Microstrip
To demonstrate the concept of the EME microstrip as a slow-wave line, the electric and magnetic surfaces are made on a two-sided printed RO4003TMcircuit board of 8mil thickness (h3) and relative dielectric constant £`
r1
of 3.38, which is
then glued to the grounded RO4003TM substrate of thickness (h2) 20mil as shown in Figure 2.4. For the EME microstrip line, copper both sides of the supporting RO4003TM substrate is etched away and kept at a distance (h1) 40mil from the ground plane. The cells size is designed of 90mil by 90mil to get the operative frequency at 10GHz. A 15-cell long EME microstrip prototype of WE = 0 mil and WM = 90mil, that is one period in the transverse direction, was fabricated and measured as shown in Figure 2.5. The EME microstrip becomes a magnetic-surface (MS) microstrip with 100% periodical structure filling the metal strip. The magnetic surface was built by applying an engraving machine followed by via through-hole plating.
We compare the measured two-port scattering parameters of the EME microstrip against those obtained by the full-wave integral equation method that assumes substrates of infinite extent. Figure 2.6 illustrates the results for S11 and S21, showing good agreement between the theoretical and measured data. The measured scattering parameters are somewhat shifted to right by approximately 0.95GHz.
21
Figure 2.5 Photograph of the 15-cell long
Careful examination on the prototype indicates the engraving machine trims more metals than required, resulting in decrease in coupled capacitance between cells and thus moving the stopband toward higher frequency. The measurements show that the stopband is between 8.25 GHz and 11.25 GHz, centered at 9.75 GHz. Inside this stopband, the reflection coefficient is nearly one (0.95) with phase of 0 degree, showing the existence of an extremely high impedance state. When operating in the lower frequency region, the magnetic-surface (MS) microstrip displays good transmission characteristics with low reflection, although it is not exactly matched to the measurement system of 50Ù impedance.
Figure 2.7 plots the phase constant of the dominant forward traveling wave component for the entire spectrum of interest. A bump appears at 8 GHz near the lower frequency corner of the stopband. Furthermore in the lower frequency region where the EH0 mode of propagation dominates, we may extract the complex propagation constants and characteristic impedance from the scattering parameters.
In lower frequency, we first compare the slow-wave factor (£f0/£fg) of the EME strip against the conventional microstrip. For example, the 100% magnetic surface (MS) microstrip shows nearly 60% increase in the SWF and 38% increase in the characteristic impedance at 5GHz. Therefore we may increase the slow-wave factor and maintain the Q-factor if we operate the EME microstrip at the
22
low-frequency region.
In the stopband, some space harmonics enter the leaky wave region and show little radiation. Therefore when a low loss slow wave line is desired, one should avoid using the EME microstrip near or above the lower corner frequency of the stopband. Analyzing the scattering parameters up to 5 GHz, the loss per guided wavelength of the EME microstrip is approximately the same as that of the conventional microstrip. Consequently, we may increase the slow-wave factor and maintain the Q-factor if we operate the EME microstrip at low-frequency region.
Figure 2.5 Photograph of the 15-cell long Electric-Magnetic-Electric (EME) microstrip.
23
S11(simu.)S11(meas.)S21(simu.)S21(meas.)0-10-20-30dB-40-50-60-7001234567101112131415Frequency (GHz)
Figure 2.6 Measured and computed scattering parameters for the EME microstrip.
24
Figure 2.7 Slow-wave factor and characteristic impedance of the MS microstrip (or EME microstrip of 100% filling) in comparison with the uniform microstrip.
25
CHAPTER 3
Size Reduction of Microstrip Patch Antennas
Using Periodic Structures
3.1 Introduction
Microstrip patch antennas are currently used for many applications due to several
advantages over wire and metallic antennas. These advantages include low cost, low profile, lightweight, and easy of integration with RF devices. However, the size becomes larger for applications at low frequency. One such a scenario is for use on a mobile communications handset. Frequency of less than 3 GHz requires conventional patches too large to be placed in mobile handset. Recently, several techniques have been proposed to minimize the microstrip patch antennas. Using high dielectric constant material to reduce size has been proposed [4], however, this method usually results in narrow bandwidth and degradation of gain. Besides, the limited availability of low loss, low cost, high dielectric constant material is another problem with this method. Figure 3.1 shows the configuration of this method. In [1-2], resistive and shorting posts were used in different agreements to reduce the overall size of microstrip patch antennas. As shown in Figure 3.2, it reduces the physical size considerable with a single shorting post [1]. Figure 3.3 shows the geometry of the rectangular microstrip antenna with a chip resistor loading [2].
26
Previously, photonic band-gap (PBG) materials have been proposed to improve
performances of microwave circuits and antennas. [5]. The periodic structure with slow-wave characteristic was successfully used to reduce the size of patch antennas [24]. Figure 3.4 shows the patch antenna with slow-wave structure. This slow-wave structure has been applied to a conventional patch antenna to reduce its size to 50% and have a decrease of gain. Here, we offer another technique for size reduction of microstrip patch antennas incorporating Electric-Magnetic-Electric (EME) composite metal strips. This newly developed periodic structure results in the increase of slow-wave factor (£f0/£fg), higher characteristic impedance, and the same Q-factor with a microstrip line [7]. The results show the increases of the SWF (slow-wave factor) at low frequency. Therefore the EME metal strips are utilized to minimize the patch antenna at operative frequency range.
27
Figure 3.1 Configuration of microstrip antenna of very high permittivity.
28
(a)
(b)
Figure 3.2 Schematic of probe-fed patch antennas incorporating single shorting post. (a) Circular. (b) Rectangular.
29
Figure 3.3 Geometry of rectangular microstrip antenna with chip-resistor loading.
Figure 3.4 Patch with Slow-wave (SWF) structure.
30
3.2 Patch Antenna Design
Figure 3.5(a) is the schematic of a rectangular patch microstrip antenna fed from
a microstrip transmission line. The substrate thickness t is much less than a wavelength. A rectangular patch antenna is usually operated near resonance in order to obtain a real-valued input impedance. Models are available for determining the resonance frequency with the cavity model [25]. The fringing fields act to extend the effective length of the length. Thus, the length of a half-wave patch is slightly less than a half wavelength in the dielectric substrate material. This is similar to foreshortening a half-wave dipole to achieve resonance. The amount of length reduction depends on εr, t and W. Formulas are available to estimate the resonant length [26], but empirical adjustments are often necessary in practice. An approximate value for the length of a resonant half-wavelength patch is [27]
λ εrL≈ 0.49£fd= 0.49where ë is the free-space wavelength, £fd the wavelength in the dielectric, and εr the substrate dielectric constant.
The region between the conductors acts as a half-wavelength transmission line
cavity that is open-circuited at its ends. Figure 3.5 (b) shows the electric fields associated with the standing wave mode in the dielectric. The electric field lines are
31
perpendicular to the conductors as required by boundary conditions and look much like those in a parallel plate capacitor. The fringing fields at the end are exposed to the upper half-space (z>0) and are responsible for the radiation. The standing wave mode with a half-wavelength separation between ends leads to electric fields that are of opposite phase on the left and right halves. Therefore, the total fringing fields at the edges are 180°out of phase and equal in magnitude. The model suggests an “aperture field” analysis approach where the patch has two radiating slot apertures with electric fields in the plane of the patch shown as Figure 3.5 (c). The fields along the edges associated with slot 1 and 2 are constant, whereas those along the other edges, seen in side view in Figure 3.5(b), have odd symmetry and their radiation cancels in the broadside direction and is usually neglected.
The pattern computation of a rectangular patch antenna is performed by first
creating equivalent magnetic surface current, as shown in Figure 3.5(c), from the fringe electric fields using MS = 2Ea x n, where Ea is the fringe electric field in each of the edge slots [28]. The far-field components is where
f(θ,φ)=∧
Eθ = E0cosφf(θ,φ) Eφ = -E0cosθsinφf(θ,φ) sin[βWsinθsinφ]2βWcos(sinθcosφ) βW2sinθsinφ232
The patch length L for resonance is given by 0.49£fd. The patch width W
is selected to give the proper radiation resistance at the input, often 50Ù. The principal plane patterns follow as
FE(θ) = cos(βLsinθ) 2sin[ E-plane, φ=0° FH(θ) = cos θ
βWsinθ]2 βWsinθ2H-plane, φ=90 This simple pattern expression neglects substrate effects and fringing effects.
Typical input impedances at the edge of a resonant rectangular patch range from 100 to 400Ù. An approximate expression for the input impedance (reactance is zero at resonance) of a resonant edge-fed patch is [29].
To avoid destroying the symmetry of the periodic structure we adopt a quarter-wave transformer to feed the patch antenna as shown in Figure 3.6. The antenna input impedance ZA can be matched to a transmission line of characteristic impedance Z
0
ZA = 90 εr2εr−1 (LW)2Ù (50Ù) with a section of transmission line that is a
quarter-wavelength long in the wavelength in the transmission line. This characteristic impedance of the matching section is given by
Zo1 = ZAZo
33
Figure 3.5(a)
The geometry of a rectangular patch microstrip antenna.
Figure 3.5(b) Side view showing the electric fields.
34
Figure 3.5(c) Top view of the rectangular patch antenna.
Figure 3.6 Microstrip edge feed with a quarter-wave transformer.
35
3.3 Statements of Slow-wave Microstrip Patch Antenna
Theoretical and experimental results of EME microstrips are presented and discussed in chapter 2.3, therefore we can design patch antennas with EME strips. The cells size is designed of 60mil by 60mil to make sure that the stopband locates much higher than our design frequency. A 15-cell long EME microstrip prototype is simulated as illustrated in Figure 3.7. We can observe that a stopband exists far away from our design frequency range. Consequently, we may apply this type of EME microstrips to design patch antennas by using the slow-wave character of strip line.
The geometry of the periodic structure with EME metal strips is illustrated in Figure 3.8. It displays an array of coupled inductors with via holes. The EME microstrip is metallized on a RO4003TM dielectric slab with thickness of 8mil (h1) and relative permittivity (£`r) of 3.38. Then we glue it on a grounded RO4003TM substrate of thickness 20mil (h2) and£`r equal to 3.38. Cell dimension is designed to be 60 by 60 mil2 with 8mil gap.
As mentioned in chapter 3.2, the resonant length of the rectangular microstrip antenna is about one half wavelength. Here we directly feed the patch by using a quarter-wave transformer to prevent destroying the symmetry of the slow-wave patch antenna as shown in Figure 3.9. Figure 3.9 (a) and Figure 3.9 (b) show the layout of
36
the slow-wave patch antenna and the conventional patch with quarter-wave transformer, respectively. The total resonant length and area of the slow-wave patch antenna are 1010 mil and (1010x1344) mil2. On the other hand, the conventional patch is made from the same RO4003TM substrate (thickness = 28mil and £`r = 3.38). The resonant length and area of the conventional one are 1330 mil and (1330x1796) mil2. The resonant length and area are reduced to be 76% and 57 % in comparison with the conventional patch.
37
0
-10
-20-30-40-50-60
0
2
4
6
8
10
12
14
16
18
20
S11S21
Figure 3.7 Simulated scattering parameters for the EME microstrip with 15 cells.
38
Figure 3.8 Geometry of the slow-wave patch with £`r = 3.38, h1 = 8mil, h2= 20mil.
39
(a) Slow-wave patch.
(b) Conventional patch.
(a) Slow-wave patch. (b) Conventional patch.
Figure 3.9 Microstrip edge feed with a quarter-wave transformer.
40
3.4 Measurement Results
We illustrate the measured return loss versus frequency for the two patch
antennas as Figure 3.10. The conventional patch has the minimum return loss of –22.98 dB at 2.57 GHz and bandwidth of 1%. However, the slow-wave patch antenna measured a peak return loss of –26.15 dB at 2.56 GHz and bandwidth of 1%. Both two antennas have the return losses lower than -20 dB. The experimental results exhibit that we can reduce the patch size by employing the EME metal strips without decreasing the bandwidth of microstrip antennas.
Figure 3.11 and Figure 3.12 show the measured E-plane and H-plane far field radiation patterns of both slow-wave and conventional patch antennas. The measured peak gains of the slow-wave and conventional patches are 4 dBi and 5.98 dBi, respectively. The co- and cross-polarization patterns on the E-plane and H-plane are both plotted in Figure 3.11, Figure 3.12. Higher level of the back lobes of the slow-wave patch antenna when compared with the traditional one leads to the degradation of peak gain.
When cross-polarization are observed, nevertheless, the –17.62 dBi (E-plane) and –19.71 dBi (H-plane) cross polarization levels of the slow-wave antenna are still low. As a result, the measured radiation patterns of both two patches are similar except the antenna gains.
41
-25-30
2
2.2
2.4
2.6
2.8
3
0-5-10
S11 (dB)
-15-20
Frequency (GHz)
slow-wave antennaconventional antenna
Figure 3.10 Measured return loss of the conventional and slow-wave patch antennas.
42
90
180
135-1045co-pol-20-30-40-30-20-100cross-pol225315270
slow-wave patchconventional
Figure 3.11 Measured radiation patterns for the slow-wave and conference patch antennas in E-plane.
43
90
135-1045
-20co-pol
-30
-40-30-20-100
180
cross-pol225315270
slow-wave patchconventional
Figure 3.12 Measured radiation patterns for the slow-wave and conference patch antennas in H-plane.
44
CHAPTER 4
Patch Antenna Gain Enhancement Utilizing
Periodical Structure
4.1 Introduction
Microstrip antennas are widely used in a broad range of many applications because of their advantageous features in terms of lightweight, low profile, low cost, and easy of integration with RF devices. Two technologies have been mainly purposed so far to achieve optimum performance. One is based on micromachining technology [30-31] while the other makes use of the concept of photonic bandgap substrates [17-19]. In the first case, part of the substrate underneath the radiating element is removed to realize a low effective dielectric constant environment for the microstrip antenna. By this method, power losses due to surface-wave excitation are reduced, while coupling of radiated power to space wave is enhanced. The second approach makes use of the concept of photonic bandgap substrates; this substrate is periodically loaded to create electromagnetic crystals whose surface-wave dispersion diagram presents a forbidden frequency range around the desired operative frequency. Because surface waves are suppressed, an increase of radiated power couples to space wave while increasing its gain and bandwidth.
Previously proposed PBG substrates have been employed to design a Ku-band microstrip patch antenna; this technology can enhance gain and bandwidth of
45
microstrip antennas by suppressing surface waves in the dielectric substrate. Figure 4.1 shows the schematic of the PBG patch surrounded by a square lattice of metal pads with grounding vias [6]. This PBG structure has been employed to design a Ku-band patch antenna, where an improvement in bandwidth and higher gain compared to a conventional one due to suppression of the surface waves in the dielectric substrate.
Figure 4.1 Schematic of the microstrip patch antenna surrounded by PBG lattices.
46
4.2 Conventional Patch Antenna Design
As mentioned in chapter 3.2, we can design a conventional microstrip patch antenna. An approximate expression for the input impedance (reactance is zero at
εZA = 90 r εr−12(L)W2 Ù resonance) of a resonant edge-fed patch is [23].
Here an insert feed scheme is employed to match the patch antenna to a 50Ù microstrip feed line as shown in Figure 4.2. The type of microstrip match offers the advantage of being planar and easily etched as well as providing adjustment input impedance through insert geometry changes. The input impedance is multiplied by the factor of con2(
π∆xi
) [25]. However, the inset depths may affect cross L
polarization and radiation pattern shape.
L
Figure 4.2 Microstrip edge feed with inset.
47
4.3 Periodical Structure Design
As mentioned in chapter 2, the EME microstrip will exist a stopband due to the periodical structures. In the stopband the periodical structure turns into a high-impedance state, known as a magnetic surface. Surface waves cannot propagate along the substrate, an increased amount of radiated power couples to space waves. Surface waves of microstrip patch antennas will be suppressed in the stopband. By this way, power loss due to surface waves can be reduced.
In our design, the first stopband of the periodical structure is designed at 5.5GHz with cells size of 130mil by 130mil. The periodical structures are fabricated on a RO4003TM dielectric slab with thickness of 20mil (h1) and relative permittivity (£`r) of 3.38. Then we glue it on a grounded RO4003TM substrate of thickness 8mil (h2) and£`r equal to 3.38 as shown in Figure 4.3. Figure 4.4 shows the computed scattering parameters of the design periodical structure. As a result, surface waves excited in the dielectric substrate will be suppressed, and improvement of the microstrip antenna performance will be expected.
48
d
Figure 4.3 The schematic of the microstrip antenna surrounded by EME structures with£`r = 3.38, h1 = 20mm, h2= 8mil, d=56mil.
49
0-10-20-30-40-50-60-70
1
2
3
4
5
6
7
8
9
10
S11S21
Figure 4.4 Computed scattering parameters for the design periodical structure.
50
4.4 Fabrication and Experiments
The resonant length of the microstrip antenna is about one half wavelength. We can estimate the resonant frequency by using the full-wave integral equation method. As a result, the conventional area is 650 by 650 mil2. Meanwhile, the design patch has the same size with the conventional patch, except the surrounding periodic structures.
Figure 4.5(a) and Figure 4.5(b) show the photograph of the design patch antenna and conventional patch, respectively. Three columns of periodical structure cells have been added to surround around the patch. The cell size is 130 by 130 mil2 with a via connected to the ground plane. The distance between the periodical cells and the patch antenna is roughly equal to twice the substrate thickness (d=56mil) so to avoid perturbing the TM10 cavity mode and the fringing field around the patch [32].
To confirm the design concept, a conventional microstrip antenna and the other one surrounded by periodical structures were fabricated and measured. Both two microstrip patch antennas are made on a RO4003TM circuit board of 20mil thickness (h1) and relative dielectric constant£`r equal to 3.38, which is glued to the grounded RO4003TM substrate of thickness (h2) 8mil.
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Figure 4.5 (a)
The conventional patch antenna.
Figure 4.5 (b) The patch antenna surrounded by periodical structures.
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4.5 Measurement Results
Figure 4.6 shows the measured return loss versus frequency for the two patch
antennas. The conventional patch has the minimum return loss of –29.37 dB at 4.72 GHz and bandwidth of 1.24%. However, the design patch antenna measured a peak return loss of –40.26 dB at 4.76 GHz and bandwidth of 1.45%. The experiments exhibit that the resonance frequency and the bandwidth of the design patch are slightly higher than the conventional one because the effective size is reduced. Both two antennas have the return losses lower than -20 dB. It indicates that most of the power is fed to both two patch antennas.
Figure 4.7 and Figure 4.8 show the measured E-plane and H-plane far field radiation patterns of both design and conventional patch antennas, including both co- and cross-polarization patterns. The measurements are at a frequency of 4.74 GHz, where the two antennas have the same return loss (-14.7 dB). While the patterns have been normalized here, the peak power received by the design patch is 0.75 dB higher in the H-plane and 1.03 dB higher in E-plane.
The co- and cross-polarization patterns in the E-plane and H-plane are both
plotted in Figure 4.7 and Figure 4.8. A broadside radiation is demonstrated and the cross-polarization are also observed. Nevertheless, the –18.8 dBi (H-plane) and –18.6 dBi (E-plane) cross polarization levels of the design patch are still low. The 3-dB beamwidth, measured at 4.74 GHz, are: design antenna (E-plane = 58.23o,
53
H-plane = 76.84o), conventional antenna (E-plane = 76.52 o, H-plane =76. o).
As shown in Figure 4.7, the design antenna has reduced radiation power along the dielectric substrate and smaller ripples in its E-plane, indicating an effective suppression of surface waves. The radiation along the directions è = ð/2 and è = -ð/2 is reduced by 1.82 and 1.41 dB with respect to the case of the conventional one. Also the back lobe of the design patch is slightly lower than that of the conventional patch because of the surface-wave suppression effect. Figure 4.8 indicates that the radiation pattern in the H-plane does not show significant differences between two antennas. In both the H- and E-planes, back lobes are a bit high because the radiation effect from the feed line. The problem due to the radiation from the feed line can be avoided by improving the feeding technique.
54
Measured resonance frequency, size, bandwidth, gain and 3-dB beamwidth of the two antennas are summarized in Table 4.1.
Frequency (GHz) Area (mil2) Bandwidth Peak Gain (dBi)
3dB Beamwidth (H-plane) 3dB Beamwidth (E-plane)
Design patch
4.76 650 by 650
1.45% 4.61 76.84o
58.23o
76.52 o 76. o 1.24% 3.58 650 by 650 Conventional patch
4.72
Table 4.1
Summary of two patch antennas.
55
Design patch
Conventional patch
Figure 4.6 Measured return loss of the conventional and design patch antennas.
56
13590
-10-20-3045
co-pol-40
180
-50-40-30-20-100
225315cross-pol270
design patchconventional patch
Figure 4.7 Measured normalized E-plane radiation patterns of the design and conference patch antennas.
57
180
13590
-10-20-30-40-50-40-30-2045co-pol
-100
cross-pol225315270
design patchconventional patch
Figure 4.8 Measured normalized H-plane radiation patterns of the design and conference patch antennas.
58
CHAPTER 5 Conclusions The recently developed periodic structures incorporate the concept of Electric-Magnetic-Electric (EME) composite metal strips, which can increase the slow-wave factor and have a broad stopband. Applying this new slow-wave structure, size reduction of the patch antenna are investigated and fabricated. As a result, size reduction is higher than 40% and gain decrease less than 3dB when compared with the conventional one. Gain enhancement of the patch antenna surrounded by periodic structures is also realized in the other part of this thesis.
Statements and measurement results of both microstrip antennas are all included in this thesis.
59
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